整流器論文中英文資料外文翻譯文獻(xiàn)

1、整流器論文中英文資料外文翻譯文獻(xiàn)AC Voltage and CurrentSensorlessControl ofThree -Phase PWM Rectifiers1 THREE-PHASE PWM RECTIFIERSA SystemModelingFig. 1 shows the power circuit of the three-phase PWM rectifier. The voltage equationsaregiven byeaR pL00i avaeb0R pL0i bvbec00R pL i cvc( 1)Fig. 1. Three-phase PWM rectif

2、ier without ac-side sensors.where , and are the sourcevoltage, the line current, andthe rectifier input voltage, respectively and are the input resistance and the input inductance, respectively. When the peakline voltage , angular frequency , and initial phase angle are given, assuming a balanced th

3、ree-phase system, the source phase voltage is expressedaseacos2ebE cos()3eccos(2)3( 2)Where1t0(3)A transformation matrix based on the estimated phase angle ,which transformsthree-phasevariablesinto a synchronous dq referenceframe, iscos2C3sinMcos( M32)cos( M23)Msin( M32)sin( M32)( 4)Transforming ( 1

4、) into the referenceframe using ( 4)eqcR pLM LiqcvqcedcM LR pLidcvdc( 5)where p is adifferential operatorand . MMExpressing( 5) in a vector notationeRiM LJipLi v( 6)where,eqciqcvqc0 1eivJedc ,idc ,vdc ,1 0( 7)Taking a transformation of ( 2) by using ( 4)E coseE sin( 8)WhereM( 9)Expressing ( 6) and (

5、 8)ina discrete domain, by approximating thederivative term in ( 6) by a forward difference 9, respectively,e(k 1)Ri( k1)M LJi (k 1)Li (k1)v( k1)i (k )( 10)TE cos(k1)e( k 1)E sin(k1)( 11)2Where T is the sampling period.Fig. 2. Overall control block diagram.B SystemControlThe PI controllers are used

6、to regulate the dc output voltage and the acinput current. For decoupling current control, the cross-coupling terms arecompensatedin a feed forward-typeand the source voltage is also compensatedas a disturbance. For transient responseswithout overshoot, the anti-windup technique is employed 10. Theo

7、verall control block diagram eliminating the sourcevoltage and line current sensorsis shown in Fig. 2. The estimationalgorithm of sourcevoltages andline currentsis describedin the following sections.2 PREDICTIVE CURRENT ESTIMATIONThe currentsof I a ( k) and I c (k ) can not be calculated instantly s

8、ince thecalculation time of the DSP is required To eliminate the delay effect, a stateobserver can be used In.addition, the state observer provides the filtering.effectsfor the estimatedvariable.Expressing5 in a state spaceform,( )-g12x AxBu()yCx(13)where,3R1L0ABL10R1CL ,0,01Lxiqcueqcvqcidcedcvdc,An

9、d y is the output.Transforming ( 12) and( 13)into a discretedomain, respectively,X (k1)FX ( k)GU ( k)( 14)Y (k)HX ( k)(15)where,1RT1 TT0FLGLRT1TT01L,LThen, the observerequationadding anerror correction term to is given byGU (k )K (Y (k)( 16)X (k 1)F X (k)Y (k )Where K is the observer gain matrixand

10、“ ” means the estimated1) is the statevariable estimatedaheadone sampling period.quantity,and X (kSubtracting ( 15)from( 16) , the error dynamic equation of the observer isexpressedaserr (k1) FKC err (k )(17)whereerr (k ) X (k). Here, it is assumedthat the model parametersmatchX (k )well with the re

11、al ones. Fig. 3 showsthe block diagram of the closed-loop state observer.The state variable error depends only on the initial error and is independentof the input . For ( 17) to convergeto the zero state,the rootsof the characteristicequationof ( 17) should belocatedwithin the unit circle.Fig. 3. Cl

12、osed-loop state observer.Fig. 4. Short pulse region.4 EXPERIMENTS AND DISCUSSIONSA. SystemHardware ConfigurationFig. 5 showsthe systemhardwareconfiguration. The sourcevoltage isa three-phase, 110 V. The input resistanceand inductance are 0.06and 3.3 mH, respectively. The dc link capacitance is 2350F

13、and the switching frequency of the PWM rectifier is 3.5 kHz.5Fig. 5. System hardware configuration.Fig. 6. Dc link currents and corresponding phase currents ( in sector V ) .The TMS320C31 DSP chip operating at 33.3 MHz is used as a main processorand two 12-b A/D convertersare used. One of them is de

14、dicatedfor detecting the dc link current and the other is usedfor measuringthe dc output6voltage and the sourcevoltages and currents, where ac side quantities are just measuredfor performancecomparison.One of two internal timers in the DSP is employed to decide the PWM control period and the other i

15、s usedto determinethe dc link current interrupt. Considering the rectifier blanking time of 3.5 s, A/D conversion time of 2.6 s, andthe other signal delay time, the minimum pulse width is setto 10 s.A. Experimental ResultsFig. 6 showsmeasureddc link currentsand phasecurrents. In caseof sectorV of th

16、e space vector diagram, the dc link current corresponds to for the switching state of and for that of . Fig. 7( a) shows the raw dc link current before filtering . It has a lot of ringing componentsdue to the resonanceof the leakageinductanceandthe snubbercapacitor. When the dc current is sampled at

17、 the end point of the active voltage vectors as shown in the figure, the measuringerror canbereduced.Fig. 7. Sampling of dc link currents.7Fig. 8. Estimated source voltage and current at starting.To reduce this error further, the low passfilter should be employed, of which result is shown in Fig. 7(

18、 b) . The cut-off frequency of the Butterworths second-order filter is 112 kHz and its delay time is about 2 sec. Since the ringing frequency is 258 kHz and the switching frequency is 3.5 kHz, the filtered signal without significant delay is acquired.Fig. 8 shows the estimatedsourcevoltage and curre

19、nt at starting. With the proposedinitial estimation strategy, the starting operation is well performed. Fig. 9 showsthe phaseangle,magnitude, andwaveform of the estimatedsource voltage, which coincide well with measuredones.Fig. 10 showsthe source voltage and current waveform at unity power factor.

20、Figs. With the estimated quantities for the feedback control, the control performanceis satisfactory. The dc voltage variation for load changeswill be remarkably decreasedif a feedforward control for theload current is added,which is possible without additional cur-rent sensor when the PWM rectifier

21、 iscombined with the PWM inverter for ac motor drives.8Fig. 9. Estimated source voltage in steady state.( a) phase angle ( b) magnitude ( c) waveform.Fig. 10. Source voltage and current waveforms.(a) estimated ( b) measured.4 CONCLUSIONSThis paper proposed a novel control scheme of the PWM rectifier

22、s without employing any ac input voltage and current sensorsand with using dc voltage and current sensorsonly. Reducing the number of the sensorsused decreasesthe systemcost aswell asimproves the systemreliability . The phase angle and the magnitude of the source voltage have been estimated by contr

23、olling the deviation betweenthe rectifier current and its model current to9be zero. For line current reconstruction,switching statesand measureddc link currents were used. To eliminate the effect of the calculation time delay of the microprocessor,the predictive stateobserver was used. It was shown

24、that the estimation algorithm is robust to the parametervariation. The whole algorithm has been implemented for a proto-type 1.5 kVAPWM rectifier system controlled by TMS320C31 DSP. The experimental results have verified that the proposedac sensorelimination method is feasible.無(wú)交流電動(dòng)勢(shì)、電流傳感器的三相PWM 整流器

25、控制1 三相 PWM 整流器10A 系統(tǒng)模型圖一所示為三相 PWM 整流器的主電路，電壓等式給出如下：eaR pL00i avaeb0R pL0i bvbec00R pL i cvc( 1)圖 1無(wú)交流傳感器三相PWM 整流器其中 e，i 和 v 分別是源電壓，線電流和整流器的輸入電壓，R 和 L 分別是輸入電阻和輸入電感。當(dāng)已知線電壓峰值 E，角頻率 和初始相位角 時(shí)，假定三相系統(tǒng)是平衡的，則源相位電壓可以表達(dá)為eacos2ebEcos()3eccos(2)3( 2)其中t0( 3)一種基于估計(jì)相位角m 的變換矩陣，將三相變量變換成一個(gè)同步的，d q 坐標(biāo)系，這個(gè)矩陣是cos Mcos( M

26、32)cos( M2)23C3sin(32)sin( M32)sinMM(4)將（ 1）式變?yōu)?dq 坐標(biāo)系使用式（ 4）eqcRpLM LiqcvqcedcM LRpLidcvdc(5)其中 p 是一個(gè)微分算子且MM11將（ 5）式寫成矢量形式eRiM LJipLiv其中eqciqcvqc01eivJedc ,idc ,vdc ,10用式（ 4）對(duì)（ 2）式進(jìn)行變換E coseE sin其中M通過(guò)前向差分來(lái)接近微分的限幅，分別將（6）式和（ 8）式用離散域表示e(k 1)Ri( k1)M LJi (k 1)Li (k1)v(k1)i (k )TE cos(k1)e(k 1)E sin(k1)

27、其中， T 是采樣周期( 6)( 7)( 8)( 9)( 10)( 11)圖 2 總的控制模塊圖B 系統(tǒng)控制PI 控制器是用來(lái)調(diào)節(jié)直流輸出電壓和交流輸入電流的。對(duì)于解耦電流控制，交叉耦合項(xiàng)用前饋式補(bǔ)償，同時(shí)，源電壓作為擾動(dòng)的補(bǔ)償。對(duì)于沒(méi)有過(guò)調(diào)的暫態(tài)響應(yīng)，引入 anti-windup 技術(shù)。消除源電壓和線電流傳感器的總的控制模塊圖如圖2 所示。源電壓和線電流的估計(jì)算法在以后的章節(jié)中介紹。122 預(yù)測(cè)電流估計(jì)由于 DSP 存在計(jì)算時(shí)間，所以 I a (k ) 和 I c ( k) 不能立即計(jì)算。為了消除延遲的影響，可以使用狀態(tài)監(jiān)測(cè)器。另外，狀態(tài)監(jiān)測(cè)器可以對(duì)估計(jì)變量起到濾波作用。將式（ 5）用狀態(tài)空

28、間形式表達(dá)為gx AxBu(12)yCx(13)其中R1L0ABL10R1L ,0,CL01iqceqcvqcxuidc,edcvdcY 是輸出。分別將式（ 12）和式（ 13）分別變換成離散領(lǐng)域X ( k1)FX (k )GU (k )( 14)Y(k) HX (k )( 15)其中1RT1TT0FLRGLT1T10T,LL則加入了誤差調(diào)整的監(jiān)測(cè)器等式為GU (k )K (Y(k )( )X ( k1) F X (k)Y( k)16其中， k 是監(jiān)測(cè)器增益矩陣，“是提前一個(gè)采樣周期”是指估計(jì)量， X (k 1)估計(jì)的狀態(tài)變量。用式（15）和減去式（ 16），監(jiān)測(cè)器的動(dòng)態(tài)誤差等式表述為err

29、(k 1) FKC err (k )( 17)其中 err (k ) X (k)7 所示X (k) 這里，假設(shè)模型參數(shù)與真實(shí)系統(tǒng)吻合的很好。圖是閉環(huán)狀態(tài)監(jiān)測(cè)器的模塊圖。13狀態(tài)變量誤差僅取決于初始誤差，與輸入無(wú)關(guān)。為了使式（ 17）趨于零狀態(tài)，典型等式（ 17）的根應(yīng)該限制在單位圓內(nèi)。圖 3閉環(huán)狀態(tài)監(jiān)測(cè)器圖 4 短脈沖區(qū)域3 實(shí)驗(yàn)與討論A 系統(tǒng)硬件構(gòu)造14圖 5系統(tǒng)硬件結(jié)構(gòu)圖 6 直流電流和相應(yīng)相電流(扇區(qū) 5).圖 5所示是系統(tǒng)的硬件結(jié)構(gòu)圖。源電壓是三相110V。輸入電阻和電感分別為0.06和 3.3mH。直流側(cè)電容為 2350F，PWM 整流器的開(kāi)關(guān)切換頻率為 3.5KHZ . 使用 TMS320C31